# Direct S-Band Phase Modulator for Space Applications

Parent Category: 2015 HFE

By D.V. Ramana, Sourabh Basu and Jolie. R

This paper describes the design, simulation and measured results of an S-band direct phase modulator. The phase modulator is realized based on the principle of a reflection modulator using 90° branch line hybrid couplers and varactor diodes. The diode is used in its linear region of operation based on simulations using ADS software. A tone processing circuit is also designed to generate appropriate bias to the modulator. The amplitude of the tone can be varied to tune the modulator for the desired modulation index. The circuits are fabricated, tested and qualified for space applications.

Introduction

A modulated microwave signal is required in almost every type of wireless application including space applications where the main concerns are the size and weight of the circuit. In order to meet these requirements, a direct phase modulator can be an appropriate solution.

There are two types of phase modulators – transmission type and reflection type. The requirements of a microwave phase modulator are to maintain a linear relationship between the phase and the modulating signal voltage and to have no change in the carrier amplitude with a wide band modulating signal and large modulation index. It is difficult to satisfy both amplitude and phase requirements in a transmission type modulator.

In a reflection type modulator, if devices and circuits including circulators are considered to be lossless, then the above constant amplitude requirement can be met. Thus, ideally, only the phase requirements need to be considered in a reflection type modulator [1].

If the carrier signal is phase modulated [2], the resultant signal is

E(t)=J0(m)cos(ωc)t - J1(m){cos(ωc ωm)t– cos(ωc + ωωm)t} + J2(m){(cos(ωc–2ωm)t

+cos(ωc +2ωm)t} -  …. ….., where

Jn(m) is the Bessel function of the first kind, order n

ωc  = carrier frequency

ωm = modulating signal frequency

m   = modulation index

In phase modulation, the modulation index is proportional to the amplitude of the modulating signal, independent of its frequency,

m α Vm , where

m  = Modulation index and peak phase deviation

Vm = Peak modulating signal amplitude (volts)

A reflective type phase shifter is used. For low power analog applications, a varactor diode phase shifter offers considerable potential. Variable reactance of varactor diodes under reverse bias condition is utilized in the phase shifter. Varactor diode phase shifters commonly make use of reflection type circuits, of which, the hybrid coupled type is very popular.

Figure 1 • Schematic of Reflection type Phase Modulator.

This paper details the development of two circuits, a direct S-band phase modulator and a tone processing circuit which provides the desired modulating input (tone) to the modulator. Both circuits are integrated and tested under appropriate environmental conditions. Simulation and hardware results are also presented.

S-Band Phase Modulator

A phase modulator at S-band is realized as a reflection type of modulator using 90° branch line hybrid couplers and varactor diodes (DH76022, M/S TEKELEC TEMEX).  The schematic of the reflection type phase modulator is shown in Fig.1. The RF input i.e. the S-band carrier is modulated by the applied modulating tone input signal to get a phase modulated (PM) output. The layout of the phase modulator is shown in Fig.2.

Figure 2 • Layout of Phase Modulator.

The phase modulator is realized using two stages to get desired phase shift. Each stage has a hybrid coupler with varactor diodes as shown in Fig.1. The carrier fed to the input port of the first stage hybrid coupler encounters variable reactance offered by the reverse biased varactor diodes. This causes the incident wave to get reflected with a phase shift proportional to the reactance offered by the varactor diodes, which in-turn, is proportional to the modulating signal. The reflected signals combine at the isolated port and the combined output is fed as input to the second hybrid coupler. The second coupler stage is identical to the first stage. The final phase modulated signal is obtained at the isolated port of the second stage coupler.

The four bias networks are implemented using quarter wavelength radial stubs, spaced by high impedance quarter wavelength lines, which offer an RF open circuit at the operating frequency, but ensure a DC path for the diodes.

A sinusoidal modulating tone of 3.9 MHz is fed to the varactor diodes through the bias networks. Tuning elements C1-C4 & L1–L8 are provided in the circuit to improve linearity, if required. However, in the existing configuration, acceptable linearity is achieved without the use of these elements.

Tone Processing Circuit

A TCXO at 62.5 MHz is selected to derive the tone frequency. Two stages of frequency dividers with division factors of 8 and 2 respectively are used to get a tone at 3.9 MHz. The ÷ 8 (PE9313) and ÷ 2 (PE9311) pre-scalars from M/S Peregrine are used. An LDO (3301A2) from M/S IR is used to provide +3V supply to the pre-scalars. The output of the pre-scalars is a bipolar square wave. This is fed to an operational amplifier (LM7171) from M/S National Semiconductor. The op-amp circuit is tuned to provide appropriate bias voltage swing and DC level shift to bias the varactor diodes in phase modulator. The output is filtered using a low pass filter to remove the unwanted frequency components and produce a sinusoidal tone at 3.9 MHz, which is fed as modulating signal to the phase modulator. The block diagram along with waveforms and the schematic of the tone processing circuit are shown in Figs. 3 and 4, respectively.

Figure 3 • Block Diagram of Tone Processing Circuit.

Figure 4 • Schematic of Tone Processing Circuit.

The low pass filter is designed for a cut-off frequency of slightly higher than 3.9 MHz, i.e., 4MHz, to accommodate frequency shift due to temperature. It is a lumped element filter. The schematic of the filter is shown in Fig. 5 and its simulated and measured responses compared in Fig.6.

Figure 5 • Schematic of Tone LPF.

Figure 6 • Comparison of Simulated & Measured Responses for LPF.

Phase Modulator Specifications

Input Carrier Frequency: 2296.875MHz

Modulating Tone Frequency: 3.9 MHz

Insertion loss: 5 dB (typ.)

Return loss: 14 dB (typ.)

Temp Range: - 20C to + 60C

Simulation Results

The varactor diode (DH76022) is simulated using Agilent ADS software to obtain a phase shift vs bias voltage curve, as shown in Fig. 7. The linear region of operation of the diode is identified in the figure and proposed for the phase modulator application. The proposed region is expanded in Fig.8 and it is seen that the linear region is centered at 3.75V with a voltage swing of 2.5V p-p.

Figure 7 • Phase Variation over Various Bias Points.

The phase modulator is also simulated using ADS software and optimized at the desired center frequency of 2296.875 MHz. The simulated S-parameters are shown in Fig. 9.

Figure 8. Phase Variation over Proposed Operating Swing (expanded from Fig.7).

Figure 9 • S-parameter Simulation for Phase Modulator.

Hardware Test Results

The phase modulator is characterized by applying bias voltage to the diodes. Based on the above simulation results, the bias voltage for the modulator is chosen from 2.5V to 5V. The modulator performance is studied for various bias voltages in the specified voltage range. The operating point is selected as 3.65V in the linear region. The sideband level and in turn the modulation index is varied by adjusting the voltage swing of the generated tone. The modulation index is adjusted to 0.7 rad., as per specification. The corresponding voltage swing is 1.3 V, which is within the linear limits of operation of diode, as obtained by simulation. The performance of the circuit is evaluated over temperature: - 20C (cold), + 25C (ambient) and + 60C (hot). The S-parameters for 3.65V bias at different temperatures is shown in Fig.10. The insertion loss is 4.5 to 5 dB and the return loss is 11 to 14 dB at the operating carrier frequency of 2296.875MHz.

Figure 10 • S-parameters at 3.65V Bias over Different Temperatures.

The modulator is tested at different RF input power levels (0 dBm to +12 dBm) with the selected tone and the output spectrum is observed as shown in Fig.11. The modulator loss is 4.8 dB.

Figure 11 • Spectrum of the Phase Modulated Signal.

The voltage, current, output power and modulation index for the modulator at different temperature conditions are given in Table 1. It is observed that the variation of output power and modulation index over temperature is 0.83 dB and ± 0.09 rad respectively.

Table 1 • Voltage, current, output power and modulation index.

Conclusion

This paper describes the design, simulation and hardware realization of an S-band direct phase modulator suitable for space applications. The modulator is tested for its performance over environmental conditions and the performance is satisfactory.

References

[1] C.S. Kim, Varactor S-band Direct Phase Modulator, IEEE Journal of Solid State Circuits, Vol. Sc-1, No.1, September 1966.

[2]  Mahrukh Khan, Novel S-band Direct Phase Modulator using Hybrid Coupler, IEEE-2009.